It is now common for digital signals to be transmitted over radio frequency communications channels. Received demodulated digital signals typically have AC and DC components--that is, the incoming digital bit stream is typically superimposed upon a DC level the amplitude of which depends upon several factors (e.g., the difference between the transmitter RF carrier frequency and the receiver RF tuning frequency, and the biasing of the receiver radio frequency circuitry). This DC component can vary greatly in dependence on these factors.
It usually takes on the order of 10-20 milliseconds or so for a receiver to "lock on" to the transmit carrier frequency (e.g., by operation of a tracking feedback-controlled phase locked loop local oscillator as is well known). The level of the DC component upon which the incoming digital data stream is superimposed is typically directly proportional to the difference between the transmitter carrier frequency and the frequency to which the receiver is tuned. This DC level commonly exhibits a transient maximum upon (or shortly after) initial receipt of an RF carrier by the receiver (and during the time the receiver local oscillator attempts to synchronize with the transmitter carrier frequency) before "stabilizing" to a steady state level.
To detect the digital data stream superimposed upon a DC component, digital radio receivers generally include a limiter circuit which distinguishes between the DC component and the digital signal stream superimposed upon the DC component--and produces an output responsive only to the digital signal stream. A simple such limiter circuit compares the received signal with a predetermined fixed threshold level. Any time the signal level is above the threshold level, the limiter produces a logic level 1 output--and the limiter produces a logic level 0 output whenever the level of the received signal is below the threshold level.
Since the DC level of the received signal varies with the difference between the incoming carrier frequency and the receiver tuning frequency, DC biases within the radio circuitry, and other factors, use of a fixed limiter threshold leads to signal detection errors. An improved prior art limiter averages the incoming signal in order to track changes in the receiver DC bia--and uses the signal average as the threshold level. Such averaging limiters are called "adaptive" because they adapt to changes in the incoming signal DC component.
FIG. 1A is a graphical illustration of a received digital signal waveform 10 relative to a non-adapting (i.e., fixed) DC threshold level 12. As mentioned previously, digital signal stream 10 is superimposed upon a DC level which changes with time. At a time t.sub.0, threshold level 12 can be used by a signal comparator to distinguish between digital signal logic level 0 and logic level 1 because it falls about midway between those two logic levels. As the receiver local oscillator frequency changes with respect to the transmitter carrier frequency (and/or receiver bias shift occurs), however, the DC level upon which the digital signal is superimposed shifts, causing both the digital signal logic level 0 and the digital signal logic level 1 to shift relative to fixed threshold level 12. By time t.sub.1 in the example the fixed threshold level 12 is about the same as digital signal logic level 1 and can no longer be used to differential between logic level 0 and logic level 1. As FIG. 1A demonstrates, soon after the DC component of the incoming signal shifts, a non-adapting limiter begins detecting only logic level zeros or logic level ones.
FIG. 2 is a schematic diagram of a prior art adapting limiter circuit 20. In this circuit, comparator 22 is used to compare the instantaneous value of the incoming signal V.sub.in with an integrated (averaged) version of the incoming signal (signal level averaging being performed by RC network 24). More particularly, a first resistor 26 connects V.sub.in to the non-inverting input of comparator 22, and a resistor 28 connects V.sub.in to the inverting input of the comparator. A capacitor 30 connected between the comparator inverting input and ground (reference) potential stores charge and resists changes in the voltage level present at the comparator inverting input. The voltage V.sub.c present across capacitor 30 can be used as an adapting threshold level 12, since it has a value approximately equal to the average DC level of the incoming signal V.sub.in over a short time period the duration of which depends on the time constant of RC network 24 (and is therefore about midway between signal logic level 1 and logic level 0).
When the signal level at the comparator non-inverting input exceeds the signal level at the comparator inverting input, comparator 22 detects a logic level 1 and applies the logic level 1 output signal to the input of modem 32. When the signal level at the comparator non-inverting input is less than the DC level stored by capacitor 30, comparator 22 detects a logic level 0 and applies a logic level 0 signal level to the input of modem 32.
Modem 32 groups the incoming serial digital data bits into parallel bit units (e.g., bytes) convenient for processing by microprocessor 34 and communicates these signal units to the microprocessor for analysis. The microprocessor may detect and/or decode the incoming digital signal in a conventional manner.
FIG. 1B is a graphical illustration of the same incoming digital signal 10 shown in FIG. 1A relative to an adapting threshold level 12 produced by RC network 24 of the FIG. 2 adapting limiter circuit. Because threshold level 12 adapts to the changing DC component of the incoming signal, the limiter successfully distinguishes between digital signal logic level 1 and digital logic level 0 despite changes in the DC component due to receiver bias, receiver tuning, and other effects.
Resistor 28 and capacitor 30 values must be selected appropriately to allow circuit 20 to adapt quickly enough to the changing DC component without adapting too rapidly. The time constant of RC network 24 should be sufficiently fast to adapt to drift in the incoming signal DC component. But because a long string of logic level 1s or logic level 0s changes the average DC level of the incoming signal over the duration of the string, the time constant of RC network 24 cannot be too fast or else circuit 20 begins to detect bits improperly because it "adapts" to the content of the incoming digital signal rather than only to the more slowly changing DC component on which the incoming signal is superimposed.
Further complications arise if the incoming digital signal includes significant low frequency components. For example, General Electric's Public Service Trunking Communications System transmits and receives digital data signals having significant frequency components as low as 10 Hz (e.g., subaudible digital signalling used to confirm proper channel allocation and/or to direct units to other channels in order to receive a higher priority call). The averaging process performed by RC network 24 is the equivalent of high-pass filtering in the frequency domain--and the network must pass frequencies as low as 10 Hz if modem 32 is to receive the intelligence carried by the low frequency components. The RC network must have a very slow time constant if the threshold level is not to "adapt" to the lower frequency components of the incoming digital signal.
It is highly desirable for the receiver to begin reliably detecting the incoming digital signal as soon as possible after initial carrier signal receipt. With an RC time constant which is long enough to prevent limiter circuit 20 from adapting to a 10 Hz bit rate, the circuit takes a long time to adapt to shifts in the varying DC component the incoming digital signal is superimposed upon. As a result, the receiver may take 20-30 milliseconds or more to adapt after first receipt of the carrier signal.
One possible solution to this problem is discussed in U.S. Pat. No. 4,631,737 to Davis et al (1986). This patent discloses a limiter circuit which detects the "peaks" (maxima) and "valleys" (minima) of the incoming signal. Davis et al's limiter sets a limiter threshold level to the midpoint between the average maximum signal level and the average minimum signal level. This is a general solution to the DC drift/offset problem, and also functions effectively for any type of signal (e.g., incoming digital signal sequences having more logic level 1 bits than logic level 0 bits or vice versa).
One disadvantage to this approach is that it is relatively complex--requiring minima and maxima signal level detection circuitry.
Davis et al discuss (at Column 3, lines 27-64; see also FIG. 2) a data limiter circuit within an RF receiver some circuitry of which is powered off during periods of inactivity in order to save battery power. The limiter includes a transistor which functions as a switch to change the time constant of a coupling network between the receiver and the limiter--allowing proper bias voltage level to be established on the coupling capacitor during initial receipt of a signal before the receiver has been fully activated. The transistor alters the RC time constant to a value needed for proper operation of the limiter when a battery saver circuit restores interrupted power to the receiver circuitry. Davis et al observe that this circuit suffers from the disadvantage that for FSK binary signalling the received bit stream must have nearly a fifty percent duty cycle of 1s and 0s during the time the capacitor is being rapidly charged (for otherwise the resultant bias voltge established on the capacitor is not the appropriate level needed for correct operation of the limiter).
Another approach is disclosed in U.S. Pat. No. 4,575,863 to Butcher et al (1986). Butcher's fast recovery biass circuit includes a limiter circuit which adaptively alters the receiver limiter operating bias level based on detection of a word synchronization bit pattern. The combination of resistors 24 nd 26 (see FIG. 2) forms a low-pass filter with a corner frequency of about 50 Hz when a switch 40 is activated. Capacitor 32 rapidly charges to the average voltage of the received data signal. Comparator 34 processes the received data signal to provide binary 1-0 information to the data decoder 20.
If a word sync binary pattern is detected by the Butcher data decoder 20, switch 40 is deactivated--reducing the corner frequency of the input circuit to approximately 5 Hz. The increased time constant (that is, reduced corner frequency) prevents reference voltage shifts which could normally occur due to long strings of ones and zeros in a binary signal pattern. When an end of message condition is detected by decoder 20, switch 40 is once again activated to decrease the input circuit time constant.
The Butcher et al limiter arrangement provides increased limiter adaptivity rate at the beginning of a received message. However, further improvements are possible.
In General Electric's new "Public Service Trunking" communication protocol (see commonly-assigned application Ser. No. 056,922 of Childress entitled "Trunked Radio Repeater System" filed June 3, 1987), a "slotted" dedicated control channel is used to pass digital channel allocation request signals from mobile transceivers acquiring service to the site controller. In an exemplary embodiment described in that copending application, the control channel is fully duplexed so that there may be simultaneous "in-bound" and "out-bound" control channel signalling. The system is "digitally" trunked in that trunking control is effected by digital signals passed over the continuously dedicated time division multiplexed "control" data channel.
FIG. 3 schematically shows calling protocol for in-bound and out-bound frequencies of the dedicated control channel in GE's Public Service Trunking System. A mobile transceiver desiring to communicate via a repeater transmits a channel request message on the control channel in-bound frequency--this channel request message having a total duration (including time for transmitting bit and word synchronization fields and error checking fields) of only 30 milliseconds. The repeater receives, decodes and processes the channel request message, and within 60 milliseconds after the last part of the channel request message has been received, transmits a responsive channel assignment message on the outbound control channel (this assignment message specifying a working channel frequency as well as other information). The mobile transceiver receives the channel assignment message and changes frequency to the working channel specified by the channel assignment message--the entire channel request/allocation process being completed within 280 milliseconds after the time the calling mobile transceiver began transmitting the channel request message.
FIG. 4 is a schematic diagram of exemplary formats for the messages shown in FIG. 3. Reference numeral 80 refers to the outgoing recurring signals transmitted by the repeater on the out-bound control channel frequency, while reference numeral 82 refers to the channel request message transmitted by the mobile transceiver on the in-bound control channel frequency.
In the preferred embodiment, the channel request message is preceded by 152 bits of dotting pattern (i.e., a string of alternating binary valued bits--101010). Following the dotting pattern, three repetitions of a work ("frame") synchronization code (a 16 bit Barker code in the preferred embodiment) are transmitted, after which is transmitted a 40 bit message specifying the type of communications required and an indentification of the calling and called mobile units (in the preferred embodiment, the message portion is transmitted three times, once inverted, to increase the probability of correct reception).
In the preferred embodiment, all signalling occurs at 9600 baud in order to improve system response time. One of the diagram objectives set forth in the "APCO-16 Requirements" (published by the Association of Police Communications Officers) is that any user must have voice channel access within one-half second after engaging a push-to-talk (PTT) switch. The exemplary embodiment utilizes the highest possible data rate (e.g., 9600 bps on the typical 25 KHz bandwidth radio channel) for critical control channel signalling in order to ensure very rapid voice channel access. The 320 bit-long channel request message transmitted on the in-bound control channel frequency by the mobile unit is transmitted at 9600 baud in only 33 milliseconds or so--and the initial 152 bit dotting portion of the channel request message has a duration of only about 16 milliseconds.
It will be appreciated by those skilled in the art that when such short message durations are involved, it is especially critical for receivers (e.g., the repeater receiver) to very rapidly begin properly detecting incoming signal levels. For example, the repeater receiver limiter must begin properly and reliably decoding the incoming signal by the time the Barker code portion of the channel request message is received if proper frame synchronization is to be acquired.
The limiter disclosed in the Butcher et al patent operates at a high adaptivity rate (time constant =20 milliseconds) until word sync (e.g., Barker code) has been received and successfully decoded. Decoding word sync, however, takes a substantial amount of time (a typical word synchronization bit pattern is 10 or more bits long). Moreover, the receiver must acquire bit synchronization before it can properly decode the word synchronization bit pattern. Significant problems result from waiting until after the word synchronization code has been properly detected before increasing the limiter adaptivity time constant.
Transmission of dotting pattern permits a receiver to very rapidly acquire bit synchronization. In fact, no other bit pattern allows a receiver to synchronize with the incoming bit timing as rapidly and efficiently. In addition, dotting pattern can be decoded very rapidly and successfully because it is a simple, alternating binary valued bit pattern--and an arbitrary desired number of received dotting pattern bits can be tested to determine whether the dotting pattern has been received.
It is therefore highly advantageous for every message to be preceded by a dotting pattern (see, e.g., commonly-assigned U.S. Pat. No. 4,663,765 issuing to Sutphin on May 5, 1987, disclosing a communications receiver which unmutes audio output in response to proper detection of a dotting pattern preamble).
Another important design goal is conservation of the processing capabilities of the receiver microprocessor used to decode and process incoming signals. When incoming signalling is received, the digital signal processor associated with the control channel typically must cease performing other tasks and begin processing the incoming signals. If the processor determines that the incoming signalling is intended for it and must be decoded, it typically must devote substantially all of its processing cycles to decoding and otherwise processing the incoming signalling. On the other hand, if the processor determines that the incoming signalling need not be decoded (e.g., the signalling may not match a predetermined protocol and therefore can be ignored), the processor can perform other useful tasks (e.g., maintenance functions) instead. It is therefore desirable for the processor to determine as rapidly as possible whether it must continue to decode incoming signalling--since reaching this decision at the earliest possible point provides more time for the procesor to perform other functions.
There is another advantage to deciding as early as possible whether incoming signalling must be decoded. Spurious signals occasionally resemble "proper" signals--sometimes causing the decoding processor to mistake the spurious signals for the ones it must decode. For example, the bit pattern of a predetermined frame synchronization word might be found embedded within a digitized voice signal transmission--or even in a received noise signal. "Falsing" occurs when the receiver's decoding circuitry mistakenly determines it has received the predetermined bit pattern it uses to distinguish "legitimate" transmissions from spurious signals when in fact it has received a spurious signal. Elimination or reduction of falsing is an important design objective, since falsing degrades overall system security and performance.
The dotting pattern preceding all messages transmitted in the General Electric Public Service Trunking System facilitates a solution to these problems which is easy to implement, reduces falsing rate and provides extremely rapid receiver adaptivity.
The dotting pattern (alternating binary valued 0 and 1 bits) is the optimal bit pattern for acquiring bit synchronization in the least amount of time--and this is why General Electic has chosen to begin each message with this dotting pattern.
Since all messages are preceded by the dotting pattern, the receiver microprocessor does not need to process any incoming data stream unless and until a dotting pattern has been detected. Falsing rate is decreased by requiring detection of a dotting pattern before procesing input signals--and the microprocessor is free to perform other tasks since it can ignore incoming signals not preceded by a dotting pattern. If a circuit external to the microprocessor is used for dotting pattern detection, the microprocessor can ignore incoming signals until the external circuit has successfully detected dotting pattern.
The present invention provides a limiter/detector which takes advantage of the characteristics of the dotting pattern which precedes each message in the General Electric Public Service Trunking System. The lowest frequency component present in a 9600 baud dotting pattern of alternating binary valued signals (101010...) is 4.8 KHz. Audio exists between 300 Hz and 3000 Hz--and the bandpass required by transmitted data extends from 10 Hz to 5 KHz. The present invention takes advantage of the dotting pattern frequency spectrum characteristics in order to shorten the bit synchronization acquisition time and improve incoming signal detection.
Since the lowest frequency component in the dotting pattern is 4800 Hz, the time constant of the receiver adaptive limiter can be decreased from 0.1 seconds to 0.33 microseconds (corresponding to a cut-off frequency of 3 KHz--much higher than the 50 Hz corner frequency used by Butcher et al, resulting in a corresponding more rapid adaptivity rate) during the time a dotting pattern is (or might be) received.
This decreased time constant allows the limiter provide by this invention to adapt much more rapidly to the DC component of the incoming data signal. The dotting pattern is detected 5 to 10 milliseconds earlier than in previous limiter signals since the limiter is able to track the "instantaneous" DC level through the transient experienced every time the transmitter begins to transmit. Upon successful decoding of the dotting pattern preceding the message, the limiter circuit time constant is changed to 0.1 seconds to allow lower frequency digital data signal components (e.g., those lower frequency components associated with word sync patterns such as Barker codes) to be passed by the limiter and detected.
The following are some of the advantages obtained by using the adaptive limiter provided by the present invention:
decreased falsing rate PA1 more reliable detection of word sync PA1 more rapid acquisition of bit sync PA1 better utilization of processing resources PA1 simplification of initial signal detection routines and/or circuitry PA1 more rapid adaptivity to incoming signalling